It is well known that hysteretic controlled switching regulators offer many performance advantages as well as configuration simplifications, which include fast response times for both input set point changes and output load changes, and no need for control loop frequency compensation or slope compensation for stability. However, for proper operation, most hysteretic switching regulators require a fairly large and repeatable value of voltage ripple signal at the output load or a means of sensing the inductor current in order to generate a control ramp signal, which is required to be supplied to a hysteretic comparator contained within the regulator. FIG. 1a illustrates an example of a prior art hysteretic regulator. As explained in more detail below, in this circuit, the equivalent series resistance of the output capacitor is utilized to generate the control ramp signal which is supplied to the hysteresis comparator.
Specifically, in the regulator of FIG. 1a, the inherent triangular waveform of the inductor current IL, flowing through the inductor 15 generated by the turning on and off of the high and low side switches 11, 12, which operate out of phase with one another, causes a ripple voltage (shown in FIG. 1b) to be developed across the equivalent series resistance (ESR) of the output capacitor C, 13. This ripple voltage is applied to the hysteresis comparator 16, causing the comparator 16 to turn on and off, thereby creating the switching control signal, which is coupled to high and low side switches 11, 12. As can be seen, the configuration of the switching regulator shown in FIG. 1 a requires a ripple voltage to appear on the output load 17, which is clearly undesirable and cannot be tolerated in the supply voltage of many systems.
Another problem with the switching regulator of FIG. 1a is that it is difficult to specify or even accurately predict the value of the equivalent series resistor (ESR) in the tantalum capacitors suitable for use in switching regulators. It is noted that ceramic dielectric capacitors have too small a value of ESR to be utilized, as the resulting ripple signal is too small. As a result, in such devices, a small value discrete resistor must usually be added in series with the output capacitor to have this configuration work successfully. The use of such discrete components, which are costly, is undesirable for various reasons.
FIGS. 2a and 2b illustrate additional prior art hysteretic switching regulators which attempt to solve some of the shortcomings of the switching regulator shown in FIG. 1a. Referring to FIG. 2a, in this configuration, a current sense resistor 19 is placed ahead of the load capacitor 13. In operation, the inductor current IL flowing through the small value current sense resistor RS 19 produces the required ramp control signal, which is coupled to the input of the hysteresis comparator 16. However, due to the current sense resistor 19, the DC voltage at the load 17 does not equal the DC voltage at the sampling point for generating the control signal, and therefore an error is introduced into the control signal, which causes an error in the regulated output voltage, VOUT. This error can be minimized by utilizing AC coupling and including an additional capacitor Cc 21 and resistor 22 in the circuit shown in FIG. 2a, as shown in FIG. 2b, which eliminates the DC component in VOUT due to Rs. However, such a configuration degrades the transient response of the switching regulator. Since the rate of change of current through an inductor is proportional to the voltage across it,
                    ⅆ                  I          L                            ⅆ        t              =                  V        L            L        ,a measure of inductor current IL can be obtained by integrating the inductor voltage:
      I    L    =            I      L        ⁢          ∫                                    V            L                    ⁡                      (            t            )                          ⁢                              ⅆ            t                    .                    This is typically done with an R-C low pass filter approximation to an integrator as shown in FIG. 3, which illustrates yet another prior art configuration of a hysteretic switching regulator.
In the switching regulator shown in FIG. 3, the RI—CI network 25 implements a low pass filter that effectively integrates the inductor voltage to obtain an inductor current signal that can be used for the ramp control signal, which is coupled to the hysteresis comparator 16. In addition, CC 21 and resistor 22 can be added as in the device of FIG. 2b to reduce the effect of the inherent series resistance of the inductor (rL) on VOUT, but it cannot be eliminated from the integration. While the configuration illustrated in FIG. 3 helps reduce the error in VOUT associated with the series resistance, Rs, in the switching regulator configuration of FIG. 2, as more components are required, including energy storage elements (i.e., inductors and capacitors), the switching regulator of FIG. 3 exhibits a poor transient response and becomes prohibitively costly, because the required component values cannot be easily implemented within an integrated circuit.
Another problem with all of the foregoing prior switching regulator circuits, which utilize inductor current to generate the triangular control signal for the comparator, is that the amplitude of the triangular signal varies with the magnitude of the load current and this causes changes in the regulator switching frequency which may be unacceptably large when the inductor and load current vary over a wide range. Other types of output filters have been proposed for generating the triangular control signal from the load voltage and inductor current, but they all have similar limitations as described above, as well as requiring several physically large R and C components that are not feasibly implemented in an integrated circuit.